DFE margin test methods and circuits that decouple sample and feedback timing

ABSTRACT

Described are methods and circuits for margin testing digital receivers. These methods and circuits prevent margins from collapsing in response to erroneously received data, and can thus be used in receivers that employ historical data to reduce intersymbol interference (ISI). Some embodiments allows feedback timing to be adjusted independent of the sample timing to measure the effects of some forms of phase misalignment and jitter.

CROSS REFERENCE TO RELATED APPLICATION

This application is a continuation-in-part of U.S. Non-ProvisionalUtility application Ser. No. 10/815,604, entitled “Margin Test Methodsand Circuits,” by Andrew Ho, Vladimir Stojanovic, Bruno W. Garlepp, andFred F. Chen, filed Mar. 31, 2004; which is a continuation-in-part ofU.S. Non-Provisional Utility application Ser. No. 10/441,461, entitled“Methods and Circuits for Performing Margining Tests in the Presence ofa Decision Feedback Equalizer,” by Fred F. Chen, filed May 20, 2003;which are incorporated herein by reference.

BACKGROUND

Signal distortion limits the sensitivity and bandwidth of anycommunication system. A form of distortion commonly referred to as“intersymbol interference” (ISI) is particularly problematic and ismanifested in the temporal spreading and consequent overlapping ofindividual pulses, or “symbols.” Severe ISI prevents receivers fromdistinguishing symbols and consequently disrupts the integrity ofreceived signals.

FIG. 1 (prior art) depicts a conventional receiver 100, which is usedhere to illustrate the ISI problem and a corresponding solution.Receiver 100 includes a data sampler 105 and a feedback circuit 110.Sampler 105 includes a differential amplifier 115 connected to adecision circuit 120. Decision circuit 120 periodically determines theprobable value of signal Din and, based on this determination, producesa corresponding output signal Dout.

Sampler 105 determines the probable value of signal Din by comparing theinput signal Din to a voltage reference Vref at a precise instant.Unfortunately, the effects of ISI depend partly on the transmitted datapattern, so the voltage level used to express a given logic level varieswith historical data patterns. For example, a series of logic zerosignals followed by a logic one signal produces different ISI effectsthan a series of alternating ones and zeroes. Feedback circuit 110addresses this problem using a technique known as Decision FeedbackEqualization (DFE), which produces a corrective feedback signal that isa function of received historical data patterns.

DFE feedback circuit 110 includes a shift register 125 connected to theinverting input of amplifier 115 via a resistor ladder circuit 130. Inoperation, receiver 100 receives a series of data symbols on an inputterminal Din, the non-inverting input terminal of amplifier 115. Theresulting output data Dout from sampler 105 is fed back to shiftregister 125, which stores the prior three output data bits. (As withother designations herein, Din and Dout refer to both signals and theircorresponding nodes; whether a given designation refers to a signal or anode will be clear from the context.)

Shift register 125 includes a number of delay elements, three flip-flopsD1-D3 in this example, that apply historical data bits to the referencevoltage side of the differential amplifier 115 via respective resistorsR1, R2, and R3. The value of each resistor is selected to provideappropriate weight for the expected effect of the correspondinghistorical bit. In this example, the value of resistor R3 is highrelative to the value of resistor R1 because the effect of the olderdata (D3) is assumed to be smaller than the effect of the newer data(D1). For the same reason, the resistance of resistor R2 is between theresistors R1 and R3. Receiver 100 includes a relatively simple DFEcircuit for ease of illustration: practical DFE circuits may sample moreor fewer historical data values. For a more detailed discussion of anumber of receivers and DFE circuits, see U.S. Pat. No. 6,493,394 toTamura et al., issued Dec. 10, 2002, which is incorporated herein byreference.

The importance of accurate data reception motivates receivermanufacturers to characterize carefully their system's ability totolerate ISI and other types of noise. One such test, a so-called“margin” test, explores the range of voltage and timing values for whicha given receiver will properly recover input data.

FIG. 2 depicts a fictional eye pattern 200 representing binary inputdata to a conventional receiver. Eye pattern 200 is graphed in twodimensions, voltage V and time T. The area of eye 205 represents a rangeof reference voltages and timing parameters within which the datarepresented by eye 205 will be captured. The degree to which the voltageV and time T of the sampling point can vary without introducing an erroris termed the “margin.”

FIGS. 3A through 3C depict three signal eyes 300, 305, and 310illustrating the effects of DFE on margins and margin testing. Referringfirst to FIG. 3A, eye 300 approximates the shape of eye 205 of FIG. 2and represents the margin of an illustrative receiver in the absence ofDFE. FIG. 3B represents the expanded margin of the same illustrativereceiver adapted to include DFE: the DFE reduces the receiver's ISI, andso extends the margins beyond the boundaries of eye 300. Increasing themargins advantageously reduces noise sensitivity and improves bit errorrates (BER).

In-system margin tests for a receiver are performed by monitoringreceiver output data (e.g., Dout in FIG. 1) while varying the referencevoltage and sample timing applied to the input waveform Din. Withreference to FIG. 2, such monitoring using various combinations ofvoltage and time permits detection of the boundaries of eye 205, wherethe boundaries are indicative of voltage and timing combinations forwhich the receiver is unable to correctly resolve the bits or symbols inthe input waveform Din. Such margin tests thus use detection of thereceipt of erroneous data to identify signal margins. Zerbe et al.detail a number of margin tests in “Method and Apparatus for Evaluatingand Optimizing a Signaling System,” U.S. patent application Ser. No.09/776,550, which is incorporated herein by reference.

A particular difficulty arises when determining the margins ofDFE-equipped receivers. While feeding back prior data bits increases themargin (FIG. 3B), the effect is just the opposite if the feedback datais erroneous. Erroneous feedback emphasizes the ISI and consequentlyreduces the margin, as shown in FIG. 3C. The margin of a DFE-equippedreceiver thus collapses when a margin test begins to probe the limits ofthe test signal (e.g., the boundaries of eye 205). The incompatiblerequirements of erroneous data for the margin test and correct data forthe DFE thus impede margin testing. There is therefore a need forimproved means of margin testing DFE-equipped receivers.

The need for accurate margin testing is not limited to DFE-equippedreceivers. Errors in margin testing lead integrated-circuit (IC)designers to specify relatively large margins of error, or “guardbands,” to ensure that their circuits will perform as advertised.Unfortunately, the use of overly large margins reduces performance, anobvious disadvantage in an industry where performance is paramount.There is therefore a need for ever more precise methods and circuits foraccurately characterizing the margins of high-speed integrated circuits.

BRIEF DESCRIPTION OF THE FIGURES

FIG. 1 (prior art) depicts a conventional digital receiver 100.

FIG. 2 depicts a fictional eye pattern 200 representing binary inputdata to a conventional receiver.

FIGS. 3A through 3C depict three signal eyes 300, 305, and 310illustrating the effects of DFE on margins and margin testing.

FIG. 4 depicts a communication system 400, including a conventionaltransmitter 402 connected to a DFE-equipped receiver 403 adapted inaccordance with one embodiment.

FIG. 5 depicts a DFE-equipped receiver 500 adapted in accordance with anembodiment to include improved means of margin testing.

FIG. 6 depicts a receiver 600 in accordance with another embodiment.

FIG. 7 depicts a receiver 700 in accordance with yet another embodiment.

FIG. 8 depicts an embodiment of a buffer 800, which may be used as oneof, amplifiers 745 in weighting circuit 735 of FIG. 7.

FIG. 9 depicts a receiver 900 in accordance with another embodiment.

FIG. 10A depicts a receiver 1000, a simplified version of receiver 900of FIG. 9 used to illustrate margin mapping in accordance with oneembodiment.

FIG. 10B is a diagram illustrating the relationship between each ofsamplers 1005 and 1010 of FIG. 10A and a data eye 1030.

FIG. 10C depicts a shmoo plot 1050 graphically depicting an illustrativemargin test in accordance with one embodiment.

FIG. 11 details an embodiment of shmoo circuit 1025 of FIG. 10A.

FIG. 12 details a receiver 1200 in accordance with another embodimentadapted to accommodate margin shmooing.

FIG. 13 depicts a receiver 1300 that supports error filtering inaccordance with another embodiment.

FIG. 14 is a waveform diagram illustrating that signal eye 1400 furtheropens after a DFE feedback signal settles.

FIG. 15 is a waveform diagram illustrating an effect of advancing theDFE feedback with respect to sample timing.

FIG. 16 depicts a receiver 1600 in accordance with an embodiment thatdecouples the feedback and sample timing to provide additional measuresof signal margin.

DETAILED DESCRIPTION

FIG. 4 depicts a communication system 400, including a conventionaltransmitter 402 connected to a receiver (receive circuit) 403 equippedwith Decision Feedback Equalization (DFE). In a normal operational mode,receiver 403 samples an input data stream from transmitter 402. Thesampled data provides DFE feedback to reduce intersymbol interference(ISI). In a margin-test mode, receiver 403 samples a known input datastream using ranges of sample timing and reference voltages. To preventa collapse of the margins, the DFE feedback path disregards thepotentially erroneous sampled data in favor of an identical version ofthe known input data stream. Such in-system margin tests can thereforeprobe the margin without collapsing the margin limits.

Receiver 403 conventionally includes respective data and edge samplers405 d and 405 e, a clock-and-data,recovery (CDR) circuit 410, and a DFEcircuit 415. During normal operation, receiver 403 receives a datastream (e.g., a series of data symbols) on sampler input terminal Din.Data sampler 405 d samples the data stream using a recovered clock RCKfrom CDR circuit 410 and produces the resulting sampled data stream on asampler output terminal Dout. Edge sampler 405 e samples the transitionsin the data stream using a recovered edge clock ECK from CDR circuit 410and delivers sampled edge data ED to CDR circuit 410. DFE circuit 415stores a plurality of prior data samples and uses these to condition theinput data in the manner discussed above in connection with FIG. 1. Inaddition to the conventional components, receiver 403 includes amultiplexer 420, an expected-data source 425, and some comparison logic430, in this case an exclusive OR (XOR) gate.

During normal operation, a test control signal T to multiplexer 420 isset to a logic zero to connect the output data Dout to the input of DFE415. Thus configured, receiver 403 acts as a conventional DFE-equippedreceiver, and CDR circuit 410 employs the edge data ED and output dataDout to derive sample clock signals RCK and ECK for respective data andedge samplers 405 d and 405 e. In a margin-test mode, however, selectsignal T is set to a logic one so as to convey an expected data streamfrom data source 425 to the input of DFE 415. Transmitter 402 thensupplies known test data on terminal Din while the expected data isapplied to DFE 415. The expected data is an identical, time-shiftedversion of the known data applied to input terminal Din, so DFE 415produces the correct feedback without regard to the output signal Dout.In essence, multiplexer 420 provides the feedback path with a firstinput terminal for sampled output data in the operational mode and witha second input terminal for expected data in the margin-test mode.

The repeated reference herein to “terminal” Din, as opposed to theplural form “terminals,” is for brevity. Receivers may include more thanone data-input terminal, such as those that rely upon differentialsignaling. Likewise, other clock, reference, and signal paths notedherein can be single-ended, differential, etc., as will be evident tothose of skill in the art. The preferred manner in which particular testcircuits and methods are adapted for use with a given receiver willdepend, in part, on the receiver architecture.

A voltage control signal CV on a like-named sampler input terminalalters the reference voltage used by sampler 405 to sample input data. Aclock control signal CC to CDR circuit 410 modifies the timing ofrecovered clock signal RCK. Control signals CV and CC are used in margintesting to explore the voltage and timing margins of receiver 403. Whenthe margin tests reach the margin limits, and thus introduce errors inoutput signal Dout, expected-data source 425 continues to provide thecorrect DFE feedback signal and consequently prevents the margins fromcollapsing in response to the errors. Comparison circuit 430 monitorsthe sampled-data series for errors by comparing the output data with theexpected data from expected-data source 425. In the event of a mismatch,comparison circuit 430 produces a logic one error signal ERR. Asequential storage element (not shown) captures any error signal.Receiver 403 thus facilitates margin testing of DFE-equipped receiverswithout collapsing the margin of interest.

Expected-data source 425 produces the same data as expected on inputterminal Din. Source 425 can be a register in which is previously storeda known data pattern to be provided during margin testing. Source 425might also be a register that goes through an expected sequence of data,such as a counter or a linear-feedback shift register (LFSR). Regardlessof the source, the expected data presents the expected output data,appropriately timed, to the input of the feedback circuit DFE 415.

FIG. 5 depicts a receiver circuit 500 in accordance with anotherembodiment. Receiver 500 is similar in some ways to receiver 403 of FIG.4, like-numbered elements being the same. Receiver 500 is extended toinclude a second sampler 505 that is substantially identical to, andconsequently mimics the behavior of, sampler 405. The margin tests areperformed on replica sampler 505 so that margin-testing circuitry haslittle or no impact on the performance of receiver 500 in theoperational mode.

Receiver 500 includes a multiplexer 510 connected to a shift register515. A modified clock and data recovery circuit CDR 520 controls thetiming of both samplers 505 and 405. Prior to a margin test, test signalT is set to logic zero and the storage elements within register 515 areloaded with an expected-data sequence. Then, in the test mode, testterminal T is set to logic one so that shift register 515 feeds itsoutput back to its input via multiplexer 510. To perform a margin test,sampler 505 samples input data Din. Comparison circuit 430 compares theresulting samples with the expected-data sequence provided by the firststorage element in register 515. Any difference between the data sampledby the replica sampler 505 and the expected sequence from register 515induces comparison circuit 430 to produce a logic one error signal online ERR. Clocking circuitry, e.g. within CDR 520, can be adapted tocontrol separately the recovered clock signals RCK1 and RCK2.

FIG. 6 depicts a receiver 600 in accordance with another embodiment.Receiver 600 is similar to the conventional receiver 100 of FIG. 1, butis modified to support improved margin testing.

Receiver 600 includes a sampler 602 that, like sampler 105 of FIG. 1,includes a differential amplifier 115 and a decision-circuit 120.Although not shown, sampler 602 includes conventional means of adjustingthe reference voltage and timing to support margin testing. DFE ofreceiver 600 performs conventionally in the operational mode andprovides expected data in the margin-test mode.

Receiver 600 includes a multiplexer 605, a comparison circuit 610, and adual-mode register 615. Multiplexer 605 conveys output signal Dout toregister 615 in the operational mode. Thus configured, receiver 600functions analogously to receiver 100 of FIG. 1. That is, register 615shifts in the output data Dout and employs three bits of historic datato provide ISI-minimizing feedback to sampler 602.

During margin testing, test signal T is set to logic one. In that case,multiplexer 605 provides the output of an XOR gate 620 to the input ofregister 615. The inclusion of XOR gate 620 and the path throughmultiplexer 605 converts register 615 into a linear-feedback shiftregister (LFSR) that provides a pseudo-random but deterministic sequenceof bits to both the input of register 615 and comparison circuit 610.Also during the margin test, the same pseudo-random sequence produced byregister 615 is provided on input terminal Din. This test sequence isapplied one clock cycle ahead of the expected data in flip-flop D1 ofregister 615, so the DFE will reflect the appropriate data regardless ofwhether output data Dout is correct. The timing and reference voltage ofsampler 602 can therefore be adjusted while monitoring output data Doutfor errors without fear of collapsing the margin limits. Comparisoncircuit 610, an exclusive OR gate in this example, flags any mismatchesbetween the output data and the expected data to identify errors.

In the example of FIG. 6, the pseudo-random sequence of test bitsapplied to input terminal Din is assumed to come from an externalsource, such as a conventional tester. The disclosed embodiments canalso be adapted to support built-in self test (BIST) or in-systemtesting. For example, a linked transmitter/receiver pair adapted inaccordance with one embodiment can margin test the intervening link. Inother embodiments, receiver 600 is modified so that register 615 oranother on-chip source provides the input test sequence. In someembodiments, register 615 is extended to include additional storageelements to produce more complex pseudo-random bit sequences. In suchcases, the number of outputs from register 615 to the input of sampler602 can be the same as or different from the number of storage elementsemployed by the LFSR. For additional details regarding LFSRs, see“What's an LFSR,” document no. SCTA036A from Texas Instruments™(December 1996) and the Xilinx™ application note entitled “EfficientShift Registers, LFSR Counters, and Long Pseudo-Random SequenceGenerators,” by Peter Alfke, XAPP 052, 7 Jul. 1996 (Version 1.1), bothof which are incorporated herein by reference.

FIG. 7 depicts a receiver 700 in accordance with yet another embodiment.FIG. 7 includes a number of elements that are incidental to theinventive margin-testing circuitry, and so are only touched upon brieflyhere. The main components of the margin-testing circuitry arehighlighted using bold outlines to distinguish them from incidentalfeatures. The emphasized components include a pair of conventionalsamplers 705 and 710 receiving input data on the same input terminal,Din, a pair of multiplexers 715 and 720, a pair of shift registers 725and 730, and a data-weighting circuit 735.

In the operational mode, multiplexers 715 and 720 both select their zeroinput. The input data Din captured by samplers 705 and 710 is thusconveyed to respective shift registers 725 and 730. The data in shiftregister 730 is the output data DATA of receiver 700, and is fed back toweighting circuit 735. For equalization feedback, all or a subset of thebits stored in the plurality of storage elements that make up shiftregister 730 are provided to weighting circuit 735. In one embodiment,shift registers 725 and 730 each store twenty bits. Of these, five bitsfrom register 730 are conveyed to weighting circuit 735. The selectedbits and their associated weighting are optimized for a given receiver.For a detailed discussion of methods and circuits for performing suchoptimization, see U.S. application Ser. No. 10/195,129 entitled“Selectable-Tap Equalizer,” by Zerbe et al., filed Jul. 12, 2002, whichis incorporated herein by reference. The details of that referencepertain to the optimization of a number of novel receivers. Themargining methods and circuits disclosed herein may be of use in anysystems that employ historical data to reduce ISI.

Weighting circuit 735 produces a weighted sum of a plurality ofhistorical bits and applies this sum to input terminal Din. This is thesame general function provided by the DFE ladder circuit of FIG. 1,though the manner in which these weighting circuits perform thisfunction differs significantly.

Weighting circuit 735 includes five amplifiers 745[0:4], each of whichreceives a bit from shift register 730. A weight-reference circuit 750provides each amplifier 745 with a reference signal (e.g., a constantcurrent) that determines the weight given to the associated bit. Theoutput terminals of amplifiers 745[0:4] are connected to input terminalDin to provide a weighted sum of five historical data values from shiftregister 730. A current-controlled embodiment of an amplifier 745[i] isdetailed below in connection with FIG. 8.

In the margin-test mode, each of multiplexers 715 and 720 selects its“one” input. The output of sampler 705 is thus conveyed to shiftregister 730 and the output of sampler 710 is conveyed to shift register725. Recall that a function of the margin-test mode is to provideexpected data to the input of the DFE circuitry. In this case, theexpected data is the input data sampled by sampler 705 and captured inshift register 730. A voltage-control signal CV2 and timing controlsignal CT2 allow a tester or test personnel to alter the referencevoltage and received clock RCK2 as necessary to probe the marginboundaries for sampler 710. Similar control signals CV1 and CT1 affordsimilar control over sampler 705 and are set to appropriate levels toensure sampler 705 correctly captures the input data.

During a margin test, potentially erroneous data bits from sampler 710pass through shift register 725. Comparison circuit 755 thereforeproduces a logic-one error signal on line ERR. In this embodiment, it isnot necessary to store expected data in advance or to provide adedicated source of expected data. Instead, the expected data is derivedfrom input data on terminal Din sampled by sampler 705. The sampler usedto produce output data in the operational mode, sampler 710, may be thesame type of sampler subjected to the margin test. Testing the receivecircuitry, as opposed to a replica, is advantageous because it providesa more accurate reading of the actual receive-circuitry performance.Also important, sampler 705 can be margined in a normal operating mode,assuming that it has independent timing and voltage control relative tosampler 710. Sampler 705 can also be margin tested and the respectivesample point (voltage and timing) centered in the data eye prior tomargin testing sampler 710.

Receiver 700 of FIG. 7 is an equalizing receiver that generates receiveand equalization clock signals. The following discussion outlinesvarious features of receiver 700. For a more detailed discussion ofsimilar receivers, see the above-incorporated application to Zerbe etal.

In addition to the components discussed above in relation to themargin-testing methods and circuits, receiver 700 includes a CDR circuit756 and an equalizer clock generator 759. Samplers 705 and 710 sampleincoming data signal Din in response to respective receive-clock signalsRCK1 and RCK2, both of which are derived from a reference clock RCLK.The samples taken by sampler 710 are shifted into register 730, wherethey are stored for parallel output via output bus DATA to someapplication logic (not shown) and to CDR circuit 756.

Receive clock signal RCLK may include multiple component clock signals,such as a data clock signal (and in some double data rateimplementations, the complement data clock signal, for capturing evenand odd phase data samples), and an edge clock signal (and optionally acomplement edge clock signal) for capturing edge samples (i.e.,transitions of the data signal between successive data eyes). The dataand edge samples are shifted into shift registers 725 and 730. Samplesin register 730 are then supplied as parallel words (i.e., a data wordand an edge word) to a phase control circuit 761 within CDR circuit 756.Phase control circuit 761 compares adjacent data samples (i.e.,successively received data samples) within a data word to determine whendata signal transitions have taken place, then compares an interveningedge sample with the preceding data sample (or succeeding data sample)to determine whether the edge sample matches the preceding data sampleor succeeding data sample. If the edge sample matches the data samplethat precedes the data signal transition, then the edge clock is deemedto be early relative to the data signal transition. Conversely, if theedge sample matches the data sample that succeeds the data signaltransition, then the edge clock is deemed to be late relative to thedata signal transition. Depending on whether a majority of suchearly/late determinations indicate an early or late edge clock (i.e.,there are multiple such determinations due to the fact that each edgeword/data word pair includes a sequence of edge and data samples), phasecontrol circuit 761 asserts an up signal (UP) or down signal (DN). Ifthere is no early/late majority, neither the up signal nor the downsignal is asserted.

Each of a pair of mix logic circuits 763 and 765 receives a set of phasevectors 767 (i.e., clock signals) from a reference loop circuit 769 andrespective timing control signals CT1 and CT2 as noted above. The phasevectors have incrementally offset phase angles within a cycle of areference clock signal. For example, in one embodiment the referenceloop outputs a set of eight phase vectors that are offset from oneanother by 45 degrees (i.e., choosing an arbitrary one of the phasevectors to have a zero degree angle, the remaining seven phase vectorshave phase angles of 45, 90, 135, 180, 225, 270, and 315 degrees). Mixlogic circuits 763 and 765 maintain respective phase count values, eachof which includes a vector-select component to select a phase-adjacentpair of the phase vectors (i.e., phase vectors that bound a phase angleequal to 360°/N, where N is the total number of phase vectors), and aninterpolation component (INT). The interpolation component INT and apair of phase vectors V1 and V2 are conveyed from each of mix logiccircuits 763 and 765 to respective receive-clock mixer circuits 770 and772. Mixer circuits 770 and 772 mix their respective pairs of phasevectors according to the interpolation component INT to generatecomplementary edge clock signals and complementary data clock signalsthat collectively constitute first and second receive-clock signals RCK1and RCK2, which serve as input clocks for samplers 705 and 710,respectively. Timing control signals CT1 and CT2 facilitate independentcontrol of the timing of clock signals RCK1 and RCK2.

Mix logic circuit 765 increments and decrements the phase count value inresponse to assertion of the up and down signals, respectively, therebyshifting the interpolation of the selected pair of phase vectors (or, ifa phase vector boundary is crossed, selecting a new pair of phasevectors) to retard or advance incrementally the phase of the receiveclock signal. For example, when the phase control logic 761 determinesthat the edge clock leads the data transition and asserts the up signal,mix logic 765 increments the phase count, thereby incrementing theinterpolation component INT of the count and causing mixer 772 toincrementally increase the phase offset (retard the phase) ofreceive-clock signal RCK1. At some point, the phase control signaloutput begins to dither between assertion of the up signal and the downsignal, indicating that edge clock components of the receive clocksignal have become phase aligned with the edges in the incoming datasignal. Mix logic 763 and mixer 770 are analogous to mix logic 765 and772, but control the receive clock RCK1 to sampler 705. These redundantcircuits are provided so the receive-clock timing to samplers 705 and710 can be independently adjusted during margin testing.

The equalizer clock generator 759 receives the phase vectors 767 fromthe reference loop 769 and includes mix logic 774 and an equalizer clockmixer 776, which collectively operate in the manner described above inconnection with mix logic 765 and mixer 772. That is, mix logic 774maintains a phase count value that is incrementally adjusted up or downin response to the up and down signals from the phase control circuit761. The mix logic selects a phase-adjacent pair of phase vectors 767based on a vector select component of the phase count. The mix logicthen outputs the selected vectors (V1, V2) and interpolation componentof the phase count (INT) to the equalizer clock mixer 776. Clock mixer776 mixes the selected vectors in accordance with the interpolationcomponent of the phase count to generate the equalizer clock signalEQCLK. The equalizer clock signal, which may include complementarycomponent clock signals, is provided to weighting circuit 735 (oranother type of equalization circuit) to time the output of equalizingsignals onto data input terminal Din.

FIG. 8 depicts an embodiment of a buffer 800 that may be used as one ofamplifiers 745 in weighting circuit 735 of FIG. 7 in an embodiment inwhich the data input Din is a two-terminal port receiving differentialinput signals Din and /Din. Clock signal EQCLK is also a differentialsignal EQCLK and /EQCLK in this embodiment.

Buffer 800 receives one of five differential feedback signals (EQDin[i]and /EQDin[i]) and the differential clock signal (EQCLK and /EQCLK) frommixer 776. Reference circuit 750 provides a reference voltage EQWi thatdetermines the current through buffer 800, and consequently the relativeweight of the selected feedback data bit.

The above-described embodiments are adapted for use in receivers ofvarious types. The embodiment of FIG. 6, for example, is applied to areceiver adapted to receive single-ended input signals, while theembodiments of FIGS. 7 and 8 are applied to receivers adapted to receivecomplementary signals. These examples are not limiting, as these andother embodiments can be applied to receivers adapted to communicatesignals in any of a number of communication schemes, includingpulse-amplitude modulated (PAM) signals (e.g., 2-PAM and 4-PAM), whichmay be used in some embodiments to provide increased data rates.

FIG. 9 depicts a receiver 900 in accordance with another embodiment.Receiver 900 is similar to receiver 700 of FIG. 7, like-identifiedelements being the same or similar. Receiver 900 differs from receiver700 in that receiver 900 omits multiplexer 715 and shift register 725.XOR gate 755 detects errors by comparing the data symbols from samplers705 and 710. As in receiver 700, both samplers 705 and 710 can bemargined in a normal operating mode. The operation of receiver 900 isotherwise similar to that of receiver 700.

Receivers 700 and 900, detailed in connection with respective FIGS. 7and 9, do not require a predetermined pattern of data (i.e., an“expected” data pattern”), and can thus be margined in the presence ofthe data patterns received during normal operation. The ability todetect system margins in system and without disrupting the normal flowof data enables accurate in-system margin test. In addition, receiversso equipped can be adapted to dynamically alter system parameters tomaintain adequate margins.

Margin Mapping (Shmoo Plots)

FIG. 10A depicts a receiver 1000, a simplified version of receiver 900of FIG. 9 used to illustrate margin mapping in accordance with oneembodiment. Receiver 1000 includes two samplers 1005 and 1010, an XORgate 1015, and a “shmoo” circuit 1025. As used herein, a shmoo circuitis used to develop shmoo data, shmoo data is information that representsmargin test results for a given sample point, and a shmoo plot is agraph that represents shmoo data to illustrate how a particular margintest or series of margin tests passes or fails in response to changes inthe reference voltage and reference timing. Samplers 1005 and 1010receive the same input data Din, but have independently adjustablereference voltages RefA and RefB and reference clocks ClkA and ClkB.

FIG. 10B is a diagram 1026 illustrating the relationship between each ofsamplers 1005 and 1010 and a data eye 1030. Each Cartesian coordinate ondiagram 1026 represents a sample coordinate, the vertical (Y) axis beingrepresentative of sample voltage and the horizontal (X) axis beingrepresentative of sample time. A data point 1035 is centered in data eye1030 along both axes, and thus represents an ideal sample point forsampler 1005.

To perform a margin test, reference voltage RefB and reference clockClkB are adjusted along their respective Y and X axes to sample datasymbols at each coordinate one or more times to probe the boundaries ofeye 1030. Margins are detected when XOR gate 1015 produces a logic one,indicating that sampler 1010 produced different data than sampler 1005.Shmoo circuit 1025 correlates errors with the respective referencevoltage RefB and clock signal ClkB for sampler 1010 and stores theresulting X-Y coordinates. Care should be taken to ensure properclock-domain crossing of the two reference clocks ClkA and ClkB toprevent data samplers 1005 and 1010 from sampling different data eyes(e.g., to prevent respective samplers from sampling different ones oftwo successive data symbols). Signals RefB and ClkB can be interchangedwith respective signals RefA and ClkA in FIG. 10B to margin sampler1010. Methods and circuits for adjusting clock phases and referencevoltages are well known in the art, and are therefore omitted here forbrevity.

FIG. 10C depicts a shmoo plot 1050 graphically depicting an illustrativemargin test in accordance with one embodiment. During margin test,reference voltage RefB and reference clock ClkB are adjusted to sampleincoming data at each voltage/time square (sample point) represented inFIG. 10C. The number of errors encountered over a fixed time is thenrecorded for each sample coordinate. The resulting plot for a givenreceiver will bear a resemblance to plot 1050, though will typically beless uniform than this illustration.

Plot 1050 can be used in a number of ways. Returning to FIG. 10B, forexample, data point 1035 is depicted in the center of eye 1030, an idealcircumstance. Plot 1050 can be used to precisely locate the true centerof eye 1030. Once this center is known, reference voltage RefA andreference clock ClkA can be adjusted as needed to maximize the marginsfor sampler 1005.

Plot 1050 can also be used to establish different margins depending uponthe allowable bit-error rate (BER) for the communication channel ofinterest. Different communication schemes afford different levels oferror tolerance. Communications channels can therefore be optimizedusing margin data gathered in the manner depicted in FIG. 10C. Forexample, an error-intolerant communication scheme might require thezero-error margin, whereas a more tolerant scheme might be afforded thelarger margin associated with a small number of errors per unit time.

Adaptive Margzining

Some embodiments detect and maintain margins without storing the shmoodata graphically depicted in FIG. 10C. One or more additional samplerscan be used to probe the margins periodically or dynamically, and thesampler used to obtain the sampled data can be adjusted accordingly. Inone embodiment, for example, the reference voltage and clock of thesampler used to obtain the sampled data are adjusted in response toperceived errors to maintain maximum margins. With reference to FIG.10A, sampler 1010 can periodically probe the high and low voltagemargins and then set reference voltage RefA between them. With referencevoltage RefA thus centered, the process can be repeated, this timeadjusting the phase of reference clock ClkB to detect the timingmargins. The phase of reference clock ClkA can then be aligned in eye1030. In other embodiments, additional samplers can simultaneously probedifferent margins of eye 1030. Dynamic margining systems in accordancewith these embodiments thus automatically account for time-variantsystem parameters (e.g., temperature and supply-voltage).

FIG. 11 details an embodiment of shmoo circuit 1025 of FIG. 10A. Shmoocircuit 1025 includes a pair of flip-flops 1100 and 1105. Flip-flop 1100synchronizes error signal Err with a clock signal Clk. Flip-flop 1105, aones detector, produces a logic-one output signal OUT in response to anylogic ones received from flip-flop 1100. In operation, both flip-flopsare reset to zero and error signal Err is monitored for a desired numberof data samples at a given timing/voltage setting. Flip-flop 1100captures any logic-one error signals Err, and ones detector 1105transitions to logic one and remains there in response to any logic onesfrom flip-flop 1100. A logic one output signal OUT is thereforeindicative of one or more error signals received in the sample period.In other embodiments, flip-flop 1105 is replaced with a counter thatcounts the number of captured errors for a given period. The number andduration of the sample periods can be changed as desired.

FIG. 12 details a double-data-rate (DDR) receiver 1200 in accordancewith another embodiment adapted to accommodate margin shmooing. Receiver1200 includes four data samplers 1205-1208 timed to an odd-phase clockClk_O, four respective flip-flops 1210 timed to an even-phase clockClk_E, three error-detecting XOR gates 1215, a multiplexer 1220,error-capturing logic 1225, and shmoo control logic 1230. An externaltester (not shown) issues test instructions and receives margin-testresults via a test-access port TAP. In another embodiment, the outputsfrom the three flip-flops 1210 following samplers 1205, 1206, and 1207connect directly to corresponding inputs of multiplexer 1220. A singleXOR gate on the output side of multiplexer 1220 then compares theselected sampler output signal with the output from sampler 1208.

As is conventional, DDR receivers receive data on two clock phases: anodd clock phase Clk_O and an even clock phase Clk_E. Receiver 1200represents the portion of a DDR receiver that captures incoming datausing the odd clock phase Clk_O. Signals specific to only one of theclock phases are indicated by the suffix “_E” or “_O” to designate aneven or odd phase, respectively. Samplers 1205, 1206, and 1207 areportions of the “odd” circuitry. Similar samplers are provided for theeven circuitry but are omitted here for brevity. The odd and even clockphases of a DDR high-speed serial input signal can be shmooed separatelyor in parallel.

Receiver 1200 may enter a shmoo mode at the direction of an externaltester and/or or under the control of another circuit internal orexternal to the receiver. Shmoo select signals Shm[1:0] then causemultiplexer 1220 to connect the output of one of XOR gates 1215 to theinput of error-capturing logic 1225. The following example assumesmultiplexer 1220 selects error signal Err1 to perform margin tests onsampler 1205. Margin tests for the remaining samplers 1206 and 1207 areidentical.

The external tester initiates a shmoo test cycle by issuing a risingedge on terminal Start. In response, control logic 1230 forces a signalRunning high and resets a ones detector 1235 within error-capturinglogic 1225 by asserting a reset signal RST. When signal Start goes low,control logic 1230 enables ones detector 1235 for a specified number ofdata clock cycles—the “shmoo-enable interval”—by asserting an enablesignal EN. When period-select signal PeriodSel is zero, the number ofdata clock cycles in the shmoo-enable interval is 160 (320 symbolperiods). When signal PeriodSel is one, the number of data clock cyclesin the shmoo-enable interval is 128 (256 symbol periods).

The lower-most sampler 1208, in response to control signals from theexternal tester, shmoos the margins for the sampler 1205 selected bymultiplexer 1220. The shmooing process is similar to that describedabove in connection with FIGS. 10A, 10B, and 10C. The process employedby receiver 1200 differs slightly, however, in that receiver 1200 takesadvantage of the presence of even clock Clk_E and flip-flops 1210 toretime the input signals to XOR gates 1215. Even clock Clk_E is 180degrees out of phase with respect to odd clock Clk_O. Clock signal ClkBcan therefore be varied up to 90 degrees forward or backward withrespect to odd clock Clk_O without fear of sampling different datasymbols with the selected sampler 1205 and sampler 1208.

The upper-most XOR gate 1215 produces a logic one if, during theshmoo-enable interval, one or more bits from sampler 1205 mismatches thecorresponding bit from sampler 1208. A flip-flop 1240 captures andconveys this logic one to ones detector 1235. At the end of theshmoo-enable interval, controller 1230 brings signal Running low andholds that state of signal Err_O. A logic one error signal Err_Oindicates to the tester that at least one mismatch occurred during theshmoo-enable interval, whereas a logic zero indicates the absence ofmismatches.

The shmoo interval can be repeated a number of times, each timeadjusting at least one of reference voltage RefD and clock CLKB, toprobe the margins of input data Din. A shmoo plot similar to that ofFIG. 10B can thus be developed for sampler 1205. This process can thenbe repeated for the remaining samplers.

Control logic 1230 does not interfere with the normal operation ofreceiver 1200, so shmooing can be performed for any type of input dataDin. Thus, receiver 1200 allows for the capture of real data eyes undervarious operating conditions, and can be used to perform in-systemmargin tests.

Other embodiments repeat the process a number of times for each of anarray of voltage/time data points to derive margin statistics thatrelate the probability of an error for various sample points within agiven data eye. Still other embodiments replace ones detector 1235 witha counter that issues an error sum count for each shmoo-enable interval.

In one embodiment, receiver 1200 samples four-level,pulse-amplitude-modulated (4-PAM) signals presented on terminal Din, inwhich case each of samplers 1205-1207 samples the input data symbolsusing a different reference voltage level. In general, the methods andcircuits described herein can be applied to N-PAM signaling schemes,where N is at least two. Such systems typically include N-1 samplers foreach data input node.

FIG. 13 depicts a receiver 1300 that supports error filtering inaccordance with another embodiment. Receiver 1300 is similar to receiver1000 of FIG. 10A, like-numbered elements being the same or similar.Receiver 1300 differs from receiver 1000 in that receiver 1300 includesdata filter 1305 that allows receiver 1300 to shmoo particular datapatterns. This is a benefit, as a receiver's margin may differ fordifferent data patterns, due to ISI for example. Data filter 1305 allowsreceiver 1300 to perform pattern-specific margin tests to bettercharacterize receiver performance.

Data filter 1305 includes a series of N data registers 1310 that providea sequence of data samples Dout to a pattern-matching circuit 1315. Inthis case N is three, but N may be more or fewer. Data filter 1305 alsoincludes a series of M (e.g., two) error registers 1320 that convey asequence of error samples to an input of an AND gate 1325. AND gate 1325only passes the error signals from registers 1320 if pattern-matchingcircuit 1315 asserts a error-valid signal ErrVal on the other input ofAND gate 1325. Pattern-matching circuit 1315 asserts signal ErrVal onlyif the pattern presented by registers 1310 matches some predeterminedpattern or patterns stored in pattern-matching circuit 1315. In oneembodiment external test circuitry (not shown) controls the patternsprovided by matching circuit 1315. Other embodiments support in-systemtesting with one or more patterns provided internally (e.g., on the samesemiconductor chip).

Measuring the Impact of DFE Settling Time

A number of the foregoing embodiments explore signal margins using twosamplers operating in parallel to receive the same incoming data stream.Depending upon their roles in the margining, one of the samplers can betermed the “main sampler” and the other the “roving sampler.” In theembodiment of FIG. 9, for example, sampler 705 can be used to capturethe incoming signal Din while the voltage and timing of sampler 710 isadjusted to explore the voltage and timing margins of the incomingsignal.

At high data rates the delay through sampler and DFE path may be suchthat the DFE feedback signal settles just in time to correct for ISIbefore the next sample instant. Using the example of FIG. 9, andassuming sampler 705 is the main sampler and sampler 710 is the rovingsampler, advancing the timing of the roving clock signal RCK2 withrespect to clock signal RCK1 may not alter the timing of feedback signalEQDin[0:4] or the delay through weighting circuit 735, and consequentlymay not alter the instant in time at which the DFE feedback is appliedto the input nodes of the samplers. Margin measurements based upon suchtiming changes may therefore not provide an accurate picture of symbolmargins.

FIG. 14 illustrates the point being made in the last paragraph. A signaleye 1400 opens to an extent after feedback signal EQDin[0:4] settles tothe correct value. The settling occurs prior to the sample time T1, andso eye 1400 is advantageously opened to provide additional voltagemargin prior to sampling by the main sampler 705. The delayedapplication of the feedback signal produces a relatively narrow portion1410 of eye 1400.

The timing of the roving sampler 710, provided by clock RCK2, can beadvanced with respect to fixed clock RCK1 to explore the narrow portion1410 of eye 1400. This is advantageous for some margin measurements, asthe narrow portion of the eye may be of interest. In practice, however,the timing of the DFE feedback typically advances with the timing of theRCK1. If, for example, the RCK1 signal were to be advanced to time T2,then the DFE feedback signal would settle earlier. In that case it maybe incorrect to assume that the narrow portion 1410 (observed when RCK1is at time T1) provides an accurate assessment of the margin of sampler705 at time T2. One embodiment thus allows DFE feedback timing to beindependent of the sample clock. With reference to FIG. 15, if DFEfeedback EQDin[0:4] were to settle well before the sample instant ofroving clock RCK2, then eye 1500 would open earlier than eye 1400 ofFIG. 14. Roving clock RCK2 could therefore be used to explore the earlyportion of a relatively open eye 1500 at e.g. time T2. This decouplingof the feedback and sample timing provides additional measures of signalmargin that can be used to better assess the true signal margins.

FIG. 16 depicts a receiver 1600 in accordance with an embodiment thatdecouples the feedback and sample timing to provide additional measuresof signal margin. Receiver 1600 includes first and second data samplers1605 and 1610, the data input ports of which are coupled to a commondata input terminal Din via a respective one of first and second summingamplifiers, or “summers,” 1615 and 1620. Summer 1615 is part of afeedback path for sampler 1605 that additionally includes a multiplyingdigital-to-analog converter (MDAC) 1625. Summer 1615 and MDAC 1625 aretogether a single-tap DFE, though additional taps may also be included.A tap weight signal Tap may be adjusted to alter the weight the feedbacksignal applied by MDAC 1625. Summer 1620 is part of a feedback path forsampler 1610 that additionally includes a multiplying MDAC 1630 and amultiplexer 1635. Summer 1620 and MDAC 1630 are together a single-tapDFE, though additional taps may also be included. MDAC 1630 shares tapweight signal Tap with MDAC 1625 in this embodiment.

Multiplexer 1635 applies expected, time-shifted values of the incomingsymbols to facilitate margin testing that explores the assumption thatDFE feedback may settle over a range of times with respect to sampleclock RCK1. Multiplexer 1635, an expected-data source, alternativelyapplies:

-   -   1. a sample data stream dSample from the data output port of        sampler 1605 (FBS<1:0>=11);    -   2. a predetermined test pattern (FBS<1:0>=10);    -   3. a steady-state voltage representative of a logic one        (FBS<1:0>=01); or;    -   4. a steady-stage voltage representative of a logic zero        (FBS<1:0>=00).        Multiplexer 1635 can also include additional inputs, such as to        the sample data stream dRoam output from sampler 1610.

Comparison circuit 1640 identifies errors by comparing the outputsignals from sampler 1605 and roving sampler 1610. An accumulator 1645,when enabled via the assertion of a valid signal Valid, increments foreach mismatch sensed by an XOR gate 1647 within comparison circuit 1640to provide a measure of the BER. As detailed below, some testconfigurations periodically make incorrect assumptions about the valuesof preceding data symbols. These incorrect assumptions may lead to falseerror signals, so comparison circuit 1640 includes a data filter thatprevents accumulator 1645 from incrementing in response to potentiallyfalse error signals. In this embodiment, the data filter includes aretimer 1650 and an XNOR gate 1655. Some test configurations do notrequire data filtering, so an OR gate 1657 is provided to disable thefiltering.

Assume feedback-select signal FBS<1:0> to multiplexer 1635 is set to“11” to direct the output signal from sampler 1605 to the input of MDAC1630. With reference to FIG. 14, the timing of the roving clock signalRCK2 can be varied, as can the offset voltage Voff to summer 1620, toexplore the boundaries of eye 1400. To a first approximation, the DFEfeedback through MDAC 1630 and summer 1620 will not shift in time withroving clock RCK2, so such a margin test will show the narrow portion1410 of eye 1400. Bit FBS1 is set to one in this mode, so OR gate 1657holds valid signal Valid high. Error signal Err is therefore assertedeach time data sample dSamp mismatches roaming data sample dRoam.Accumulator 1645, constantly enabled by valid signal Valid, accumulatesthe bit errors. Though not shown, accumulator 1645 is synchronized tothe incoming data.

Assume feedback-select signal FBS<1:0> to multiplexer 1635 is set to“00” to direct a steady-state logic zero to the input of MDAC 1630. Thisis akin to always assuming the prior received data symbol wasrepresentative of a logic zero, and providing appropriate DFE feedbackbased upon that assumption. Such a scenario is depicted in FIG. 15 asfeedback signal EQDin[0:4] having settled well before receipt of thecurrent symbol 1500. In that case, and assuming the prior receivedsymbol was indeed representative of a logic zero, the opening of thereceived eye occurs earlier than if the receiver had to await resolutionof the preceding symbol. The first portion of eye 1500 of FIG. 15 isthus more open than eye 1400 of FIG. 14. The timing of the roving clocksignal RCK2 and offset voltage Voff can then be varied to explore theboundaries of the widened eye 1500.

Eye 1500 is only widened when the assumption about the prior data symbolis correct; otherwise, the DFE feedback tends to degrade the incomingsignal. Comparison circuit 1640 is therefore adapted to disableaccumulator 1645 when the assumption about the preceding data symbol isincorrect. Retimer 1650 delays output data signal dSamp by one symboltime so that the output to XNOR gate 1655 is the resolved prior datasample. XNOR gate 1655 thus only outputs a logic one if the selectedsymbol from multiplexer 1635 matches the prior data symbol, and isconsequently the correct level for the DFE feedback signal. Selectsignal FBS<1:0> can be set to “01” to perform the same test based uponthe assumption that the preceding bit was a logic one.

In effect, the applied zero or one is an expected, time-shifted valuerepresentative of a corresponding prior input data symbol. The timing ofthe applied DFE feedback can thus be provided independently of either orboth of the sample clock signals RCK1 and RCK2.

Select signal FBS<1:0>can be set to “10” to provide a test pattern fromsource 1656. The test pattern might be, for example, an identical,time-shifted version of the known data applied to input terminal Din.The time-shift of the expected data can be varied with the roving clocksignal RCK2, though in other embodiments the timing of the test pattern,and consequently the applied DFE feedback for the roving sampler, can becontrolled separately from the roving and data clocks. Data filtering isnot used in the depicted embodiment when the DFE feedback for MDAC 1630is based upon sampled data dSamp or a test pattern.

Some of the foregoing embodiments employ an additional sampler to probethe margins of a given data input. Some receiver architectures alreadyinclude the requisite additional sampler, to support additionalsignaling modes, for example. Other embodiments may be adapted toinclude one or more additional “monitor” samplers.

An output of a process for designing an integrated circuit, or a portionof an integrated circuit, comprising one or more of the circuitsdescribed herein may be a computer-readable medium such as, for example,a magnetic tape or an optical or magnetic disk. The computer-readablemedium may be encoded with data structures or other informationdescribing circuitry that may be physically instantiated as anintegrated circuit or portion of an integrated circuit. Although variousformats may be used for such encoding, these data structures arecommonly written in Caltech Intermediate Format (CIF), Calma GDS IIStream Format (GDSII), or Electronic Design Interchange Format (EDIF).Those of skill in the art of integrated circuit design can develop suchdata structures from schematic diagrams of the type detailed above andthe corresponding descriptions and encode the data structures oncomputer readable medium. Those of skill in the art of integratedcircuit fabrication can use such encoded data to fabricate integratedcircuits comprising one or more of the circuits described herein.

While the present invention has been described in connection withspecific embodiments, variations of these embodiments will be obvious tothose of ordinary skill in the art. Moreover, unless otherwise defined,terminals, lines, conductors, and traces that carry a given signal fallunder the umbrella term “node.” In general, the choice of a givendescription of a circuit node is a matter of style, and is not limiting.Likewise, the term “connected” is not limiting unless otherwise defined.Some components are shown directly connected to one another while othersare shown connected via intermediate components. In each instance, themethod of interconnection establishes some desired electricalcommunication between two or more circuit nodes, or terminals. Suchcommunication may often be accomplished using a number of circuitconfigurations, as will be understood by those of skill in the art.Furthermore, only those claims specifically reciting “means for” or“step for” should be construed in the manner required under the sixthparagraph of 35 U.S.C. §112. Therefore, the spirit and scope of theappended claims should not be limited to the foregoing description.

1. A receive circuit comprising: a. a data input terminal to receive astream of input data symbols; b. a first data sampler having a firstdata input port, a first data output port, and a first clock terminal toreceive a sample clock signal; c. a second data sampler having a seconddata input port, a second data output port, and a second clock terminal;d. an expected-data source to provide at least one expected,time-shifted value representative of a corresponding at least one of theinput data symbols; and e. a decision feedback path extending from theexpected-data source to the second data input port.
 2. The receivecircuit of claim 1, wherein the time-shifted values of the input datasymbols have a first phase and the sample clock signal has a secondphase and the first and second phases are independent.
 3. The receivecircuit of claim 1, wherein the expected-data source is adapted toselect the first data output port to receive the time-shifted values ofthe input data signals.
 4. The receive circuit of claim 1, furthercomprising a comparison circuit having a first comparison-circuit inputnode coupled to the first data output port, a second comparison-circuitinput node coupled to the second data output node, and acomparison-circuit output node.
 5. The receive circuit of claim 4,wherein the first and second data samplers issue respective first andsecond sampled-data streams, and wherein the comparison circuit issuesan error signal in response to mismatches between the first and secondsampled-data streams.
 6. The receive circuit of claim 5, wherein thecomparison circuit issues an error signal in response to each mismatchbetween the first and second sampled-data streams.
 7. The receivecircuit of claim 1, further comprising a sample filter coupled to thecomparison-circuit output node, the sample filter adapted to rejectsamples for which the time-shifted value and the corresponding one ofthe input data symbols differ.
 8. The receive circuit of claim 1,wherein the second clock terminal is adapted to receive a second clocksignal, and wherein the first-mentioned and second clock signals are outof phase.
 9. The receive circuit of claim 1, wherein the time-shiftedvalue is a constant.
 10. The receive circuit of claim 1, furthercomprising a second decision feedback path extending from the first dataoutput port to the first data input port.
 11. The receive circuit ofclaim 1 instantiated on a single semiconductor chip.
 12. A methodcomprising: a. receiving a series of input symbols; b. adding aweighted, time-shifted version of the input symbols to the series ofinput symbols to develop a first series of equalized input symbols; c.sampling the first series of equalized input symbols using a first clocksignal to produce a first series of sampled symbols; d. deriving theweighted, time-shifted version of the input symbols from the firstseries of sampled symbols; e. adding expected data to the series ofinput symbols to develop a second series of equalized input symbols; andf. sampling the second series of equalized input symbols using a secondclock signal to produce a second series of sampled symbols.
 13. Themethod of claim 12, wherein the expected data does not transition withthe time-shifted version of the input symbols.
 14. The method of claim12, wherein the expected data comprises a steady-state valuerepresentative of a logic level.
 15. The method of claim 12, wherein theexpected data is out of phase with respect to the second clock signal.16. The method of claim 12, wherein the expected data is of a lowerfrequency than the series of input signals.
 17. The method of claim 12,further comprising selecting the first series of sampled signals as theexpected data.
 18. The method of claim 12, further comprising comparingthe first series of sampled symbols with the second series of sampledsymbols.
 19. The method of claim 18, further comprising excluding fromconsideration a subset of results of the comparing.
 20. The method ofclaim 12, wherein the weighted, time-shifted version of the inputsymbols are derived using a first feedback path and the expected dataare conveyed via a second feedback path.
 21. A receiver comprising: a. adata input terminal to receive a stream of input data symbols; b. afirst data sampler having a first data input port, a first data outputport, and a first clock terminal to receive a sample clock signal; c. asecond data sampler having a second data input port, a second dataoutput port, and a second clock terminal; d. means for applying aweighted, time-shifted expected value representative of a prior one ofthe input data symbols to the second data input port.
 22. The receiverof claim 21, wherein the weighted, time-shifted expected value comprisesa steady-state value.
 23. The receiver of claim 22, wherein theweighted, time-shifted expected value comprises a voltage level.
 24. Thereceiver of claim 21, further comprising means for comparing a firstoutput data stream from the first data output port with a second outputdata stream from the second data output port.
 25. The receiver of claim24, further comprising a data filter coupled to an output of the meansfor comparing.